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1 lt1513/lt1513-2 sn1513 1513fas sepic constant- or programmable-current/ constant-voltage battery charger the lt 1513 is a 500khz current mode switching regula- tor specially configured to create a constant- or program-mable-current/constant-voltage battery charger. in addition to the usual voltage feedback node, it has a current sense feedback circuit for accurately controlling output current of a flyback or sepic (single-ended primary inductance converter) topology charger. these topologies allow the current sense circuit to be ground referred and completely separated from the battery itself, simplifying battery switch- ing and system grounding problems. in addition, these topologies allow charging even when the input voltage is lower than the battery voltage. the lt1513 can also drive a ccfl royer converter with high efficiency in floating or grounded mode. maximum switch current on the lt1513 is 3a. this allows battery charging currents up to 2a for a single lithium-ion cell. accuracy of 1% in constant-voltage mode is perfect for lithium battery applications. charging current can be easily programmed for all battery types. descriptio n u charger input voltage may be higher, equal to orlower than battery voltage charges any number of cells up to 20v 1% voltage accuracy for rechargeable lithiumbatteries 100mv current sense voltage for high efficiency(lt1513) 0mv current sense voltage for easy currentprogramming (lt1513-2) battery can be directly grounded 500khz switching frequency minimizesinductor size charging current easily programmable or shut down features input voltage (v) 05 current (a) 2.42.2 2.0 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 25 lt1513 ?ta02 10 15 20 30 12v inductor = 10 m h actual programmed charging current will beindependent of input voltage if it does not exceed values shown single li-ion cell (4.1v) double li-ion cell (8.2v) 16v 20v battery voltage maximum charging current , ltc and lt are registered trademarks of linear technology corporation. charging of nicd, nimh, lead-acid or lithiumrechargeable cells precision current limited power supply constant-voltage/constant-current supply transducer excitation universal input ccfl driver applicatio n s u figure 1. sepic charger with 1.25a output current lt1513 i fb v c v in l1a* l1b* 1.25a 13 2 5 7 6 tab 4 gnd v fb lt1513 ?ta01 v sw sync and/or shutdown wall adapter input s/s c322 m f 25v c2** 4.7 m f c50.1 m f * ** ? l1a, l1b are two 10 m h windings on a common core: coiltronics ctx10-4ceramic marcon thcr40eie475z or tokin 1e475zy5u-c304 mbrd340 or mbrs340t3. mbrd340 has 5 a typical leakage, mbrs340t3 50 a typical c40.22 m f r4 39 w r1 r2 r5270 w r30.08 w c122 m f 25v 2 d1 ? charge shutdown + + typical applicatio n u downloaded from: http:///
2 lt1513/lt1513-2 sn1513 1513fas a u g w a w u w a r b s o lu t ex i t i s supply voltage ....................................................... 30v switch voltage ........................................................ 40v s/s pin voltage ....................................................... 30v fb pin voltage (transient, 10ms) ......................... 10v v fb pin current .................................................... 10ma i fb pin voltage (transient, 10ms) ......................... 10v operating junction temperature range lt1513c ............................................... 0 c to 125 c lt1513i ............................................ 40 c to 125 c short circuit ......................................... 0 c to 150 c storage temperature range ................ 65 c to 150 c lead temperature (soldering, 10 sec)................. 300 c e lectr ic al c c hara terist ics v in = 5v, v c = 0.6v, v fb = v ref , i fb = 0v, v sw and s/s pins open, unless otherwise noted. symbol parameter conditions min typ max units v ref fb reference voltage measured at fb pin 1.233 1.245 1.257 v v c = 0.8v 1.228 1.245 1.262 v fb input current v fb = v ref 300 550 na 600 na fb reference voltage line regulation 2.7v v in 25v, v c = 0.8v 0.01 0.03 %/v v iref i fb reference voltage (lt1513) measured at i fb pin 107 100 93 mv v fb = 0v, v c = 0.8v ?10 100 90 mv i fb input current v ifb = v iref (note 2) 10 25 35 m a i fb reference voltage line regulation 2.7v v in 25v, v c = 0.8v 0.01 0.05 %/v i fbvos i fb voltage offset (lt1513-2) (note 3) i vfb = 60 m a (note 4) 7.5 2.5 12.5 mv i fb input current v ifb = v iref 200 10 0 na v fb source current v iref = 10mv, v fb = 1.2v 700 300 100 m a g m error amplifier transconductance d i c = 25 m a 1100 1500 1900 m mho 700 2300 m mho error amplifier source current v fb = v ref ?150mv, v c = 1.5v 120 200 350 m a error amplifier sink current v fb = v ref + 150mv, v c = 1.5v 1400 2400 m a wu u package / o rder i for atio consult factory for military grade parts. order part number lt1513crlt1513-2cr lt1513ir lt1513-2ir t jmax = 125 c, q ja = 30 c/ w with package soldered to 0.5inch 2 copper area over backside ground plane or internalpower plane, q ja can vary from 20 c/w to > 40 c/w depending on mounting technique r package 7-lead plastic dd front view tab is gnd v in s/sv sw gndi fb fbv c 76 5 4 3 2 1 order part number lt1513-2ct7LT1513-2IT7 t jmax = 125 c, q ja = 50 c/ w, q jc = 4 c/w t7 package 7-lead to-220 v in s/sv sw gndi fb fbv c front view 76 5 4 3 2 1 downloaded from: http:///
3 lt1513/lt1513-2 sn1513 1513fas e lectr ic al c c hara terist ics v in = 5v, v c = 0.6v, v fb = v ref , i fb = 0v, v sw and s/s pins open, unless otherwise noted. symbol parameter conditions min typ max units error amplifier clamp voltage high clamp, v fb = 1v 1.70 1.95 2.30 v low clamp, v fb = 1.5v 0.25 0.40 0.52 v a v error amplifier voltage gain 500 v/ v v c pin threshold duty cycle = 0% 0.8 1 1.25 v f switching frequency 2.7v v in 25v 450 500 550 khz 0 c t j 125 c 430 500 580 khz t j < 0 c 400 580 khz maximum switch duty cycle 85 95 % switch current limit blanking time 130 260 ns bv output switch breakdown voltage 0 c t j 125 c4 0 4 7 v t j < 0 c3 5v v sat output switch on resistance i sw = 2a 0.25 0.45 w i lim switch current limit duty cycle = 50% 3.0 3.8 5.4 a duty cycle = 80% (note 1) 2.6 3.4 5.0 a d i in / d i sw supply current increase during switch on time 15 25 ma/a control voltage to switch current 4a / v transconductance minimum input voltage 2.4 2.7 v i q supply current 2.7v v in 25v 4 5.5 ma shutdown supply current 2.7v v in 25v, v s/s 0.6v, t j 3 0 c 12 30 m a t j < 0 c5 0 m a shutdown threshold 2.7v v in 25v 0.6 1.3 2 v shutdown delay 51 22 5 m s s/s pin input current 0v v s/s 5v ?0 15 m a synchronization frequency range 600 800 khz the denotes specifications which apply over the full operating temperature range.note 1: for duty cycles (dc) between 50% and 85%, minimum guaranteed switch current is given by i lim = 1.33 (2.75 ?dc). note 2: the i fb pin is servoed to its regulating state with v c = 0.8v. note 3: consult factory for grade selected parts. note 4: the i fb pin is sevoed to regulate fb to 1.245v downloaded from: http:///
4 lt1513/lt1513-2 sn1513 1513fas temperature ( c) ?0 ?0 negative feedback input current ( m a) ?0 0 0 50 75 lt1513 ?g06 ?0 ?0 ?0 ?5 25 100 125 150 charging current (a) battery voltage (v) 1210 86 4 2 0 0.4 0.8 1.2 1.6 1513 g07 2.0 0.2 0 0.6 1.0 1.4 1.8 (a) 8.4v battery i chrg = 0.5a (b) 8.4v battery i chrg = 1a (c) 4.2v battery i chrg = 1.5a v in = 12v maximum available charging current with 12v input (a) (b) (c) negative feedback input currentvs temperature output charging characteristicsshowing constant-current and constant-voltage operation temperature ( c) ?0 0 minimum synchronization voltage (v p-p ) 0.5 1.0 1.5 2.0 05 0 100 150 lt1513 ?g04 2.5 3.0 ?5 25 75 125 f sync = 700khz minimum peak-to-peaksynchronization voltage vs temperature temperature ( c) ?0 feedback input current (na) 400 500 600 150 lt1513 ?g05 300200 0 0 50 100 100 800700 ?5 25 75 125 v fb = v ref feedback input currentvs temperature typical perfor m a n ce characteristics u w switch saturation voltagevs switch current minimum input voltagevs temperature switch current limitvs duty cycle temperature ( c) ?0 1.8 input voltage (v) 2.0 2.2 2.4 2.6 05 0 100 150 lt1513 ?g03 2.8 3.0 ?5 25 75 125 duty cycle (%) 0 switch current limit (a) 2 4 61 3 5 20 40 60 80 lt1513 ?g02 100 10 0 30 50 70 90 25 c and 125 c ?5 c switch current (a) 0 switch saturation voltage (v) 0.6 0.8 1.0 3.2 lt1513 ?g01 0.4 0.2 0.5 0.7 0.9 0.3 0.1 0 0.8 1.6 2.4 4.0 2.8 0.4 1.2 2.0 3.6 100 c 150 c 25 c ?5 c downloaded from: http:///
5 lt1513/lt1513-2 sn1513 1513fas pi n fu n ctio n s uuu v c (pin 1): the compensation pin is primarily used for frequency compensation, but it can also be used for softstarting and current limiting. it is the output of the error amplifier and the input of the current comparator. peak switch current increases from 0a to 3.6a as the v c voltage varies from 1v to 1.9v. current out of the v c pin is about 200 m a when the pin is externally clamped below the internal 1.9v clamp level. loop frequency compensationis performed with a capacitor or series rc network from the v c pin directly to the ground pin (avoid ground loops). fb (pin 2): the feedback pin is used for positive output voltage sensing. the r1/r2 voltage divider connected tofb defines li-ion float voltage at full charge, or acts as a voltage limiter for nicd or nimh applications. fb is the inverting input to the voltage error amplifier. input bias current is typically 300na, so divider current is normally set to 100 m a to swamp out any output voltage errors due to bias current. the noninverting input of this amplifier istied internally to a 1.245v reference. the grounded end of the output voltage divider should be connected directly to the lt1513 ground pin (avoid ground loops). i fb (pin 3): the current feedback pin is used to sense charging current. it is the input to a current sense amplifierthat controls charging current when the battery voltage is below a programmed limit. during constant-current operation, the lt1513 i fb pin regulates at 100mv. input resistance of this pin is 5k w , so filter resistance (r4, figure 1) should be less than 50 w . the 39 w , 0.22 m f filter shown in figure 1 is used to convert the pulsating currentin the sense resistor to a smooth dc current feedback signal. the lt1513-2 i fb pin regulates at 0mv to provide programmable current limit. the current through r5,figure 5, is balanced by the current through r4, program- ming the maximum voltage across r3. gnd (pin 4): the ground pin is common to both control circuitry and switch current. v c , fb and s/s signals must be kelvin and connected as close as possible to this pin.the tab of the r package should also be connected to the power ground. v sw (pin 5): the switch pin is the collector of the power switch, carrying up to 3a of current with fast rise and falltimes. keep the traces on this pin as short as possible to minimize radiation and voltage spikes. in particular, the path in figure 1 which includes sw to c2, d1, c1 and around to the lt1513 ground pin should be as short as possible to minimize voltage spikes at switch turn-off. s/s (pin 6): this pin can be used for shutdown and/or synchronization. it is logic level compatible, but can betied to v in if desired. it defaults to a high on state when floated. a logic low state will shut down the charger to amicropower state. driving the s/s pin with a continuous logic signal of 600khz to 800khz will synchronize switch- ing frequency to the external signal. shutdown is avoided in this mode with an internal timer. v in (pin 7): the input supply pin should be bypassed with a low esr capacitor located right next to the ic chip. thegrounded end of the capacitor must be connected directly to the ground plane to which the tab is connected. tab: the tab on the surface mount r package is electri- cally connected to the ground pin, but a low inductanceconnection must be made to both the tab and the pin for proper circuit operation. see suggested pc layout in figure 4. downloaded from: http:///
6 lt1513/lt1513-2 sn1513 1513fas block diagra m w + i fba i fb s/s v fb 4k 50k* *remove on lt1513-2 0.04 w + ea v c v in lt1513 ?bd 1.245v ref 500khz osc sync shutdown delay and reset low dropout 2.3v reg antisat logic driver sw switch + ia a v ? 6 comp the lt1513 is a current mode switcher. this means thatswitch duty cycle is directly controlled by switch current rather than by output voltage or current. referring to the block diagram, the switch is turned ?n?at the start of each oscillator cycle. it is turned ?ff?when switch current reaches a predetermined level. control of output voltage and current is obtained by using the output of a dual feedback voltage sensing error amplifier to set switch current trip level. this technique has the advantage of simplified loop frequency compensation. a low dropout internal regulator provides a 2.3v supply for all internal circuitry on the lt1513. this low dropout design allows input voltage to vary from 2.7v to 25v. a 500khz oscillator is the basic clock for all internal timing. it turns ?n?the output switch via the logic and driver circuitry. special adaptive antisat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. this minimizes driver dissipation and provides very rapid turn-off of the switch. a unique error amplifier design has two inverting inputs which allow for sensing both output voltage and current. a 1.245v bandgap reference biases the noninverting input.the first inverting input of the error amplifier is brought out for positive output voltage sensing. the second inverting input is driven by a ?urrent?amplifier which is sensing output current via an external current sense resistor. the current amplifier is set to a fixed gain of 12.5 which provides a 100mv current limit sense voltage. the lt1513-2 option removes the feedback resistors around the i fb amplifier and connects its output to the fb signal. this provides a ground referenced current sensevoltage suitable for external current programming and makes amplifier input and output available for external loop compensation. the error signal developed at the amplifier output is brought out externally and is used for frequency compen- sation. during normal regulator operation this pin sits at a voltage between 1v (low output current) and 1.9v (high output current). switch duty cycle goes to zero if the v c pin is pulled below the v c pin threshold, placing the lt1513 in an idle mode. operatio n u figure 2 downloaded from: http:///
7 lt1513/lt1513-2 sn1513 1513fas applicatio n s i n for m atio n wu u u the lt1513 is an ic battery charger chip specifically opti-mized to use the sepic converter topology. a complete charger schematic is shown in figure 1. the sepic topology has unique advantages for battery charging. it will operate with input voltages above, equal to or below the battery voltage, has no path for battery discharge when turned off, and eliminates the snubber losses of flyback designs. it also has a current sense point that is ground referred and need not be connected directly to the battery. the two inductors shown are actually just two identical windings on one inductor core, although two separate inductors can be used. a current sense voltage is generated with respect to ground across r3 in figure 1. the average current through r3 is always identical to the current delivered to the battery. the lt1513 current limit loop will servo the voltage across r3 to 100mv when the battery voltage is below the voltage limit set by the output divider r1/r2. constant-current charging is therefore set at 100mv/r3. r4 and c4 filter the current signal to deliver a smooth feedback voltage to the i fb pin. r1 and r2 form a divider for battery voltage sensing andset the battery float voltage. the suggested value for r2 is 12.4k. r1 is calculated from: r rv ra bat 1 2 1 245 1 245 2 0 3 = +m (C .) .( . ) v bat = battery float voltage 0.3 m a = typical fb pin bias current a value of 12.4k for r2 sets divider current at 100 m a. this is a constant drain on the battery when power to the charger isoff. if this drain is too high, r2 can be increased to 41.2k, reducing divider current to 30 m a. this introduces an addi- tional uncorrectable error to the constant voltage float modeof about 0.5% as calculated by: v error = 0.15 a(r1)(r2) 1.245(r1+ r2) bat m 0.15 m a = expected variation in fb bias current around the nominal 0.3 m a typical value. with r2 = 41.2k and r1 = 228k, (v bat = 8.2v), the error due to variations in bias current would be 0.42%. a second option is to disconnect the divider when chargerpower is off. this can be done with a small nfet as shown in figure 3. d2, c6 and r6 form a peak detector to drive the gateof the fet to about the same as the battery voltage. if power is turned off, the gate will drop to 0v and the only drain on the battery will be the reverse leakage of the catch diode d1. see diode selection for a discussion of diode leakage. lt1513 v in l1a l1b gnd v fb 1513 f03 v sw adapter input c2 schematic simplified for clarityd2 = 1n914, 1n4148 or equivalent c6470pf r6470k r3 r1 r2 d2 d1 c1 + figure 3. eliminating divider current maximum input voltagemaximum input voltage for the lt1513 is partly determined by battery voltage. a sepic converter has a maximum switch voltage equal to input voltage plus output voltage. the lt1513 has a maximum input voltage of 30v and a maximum switch voltage of 40v, so this limits maximum input voltage to 30v, or 40v ?v bat , whichever is less. shutdown and synchronizationthe dual function s/s pin provides easy shutdown and synchronization. it is logic level compatible and can be pulled high or left floating for normal operation. a logic low on the s/s pin activates shutdown, reducing input supply current to 12 m a. to synchronize switching, drive the s/s pin between 600khz and 800khz.inductor selection l1a and l1b are normally just two identical windings on one core, although two separate inductors can be used. a typical value is 10 m h, which gives about 0.5a peak-to-peak induc- tor current. lower values will give higher ripple current,which reduces maximum charging current. 5 m h can be used if charging currents are at least 20% lower than the values downloaded from: http:///
8 lt1513/lt1513-2 sn1513 1513fas applicatio n s i n for m atio n wu u u shown in the maximum charging current graph. higherinductance values give slightly higher maximum charging current, but are larger and more expensive. a low loss toroid core such as kool m m , molypermalloy or metglas is recommended. series resistance should be less than 0.04 w for each winding. ?pen core?inductors, such as rods orbarrels are not recommended because they generate large magnetic fields which may interfere with other electronics close to the charger. input capacitor the sepic topology has relatively low input ripple current compared to other topologies and higher harmonics are especially low. rms ripple current in the input capacitor is less than 0.25a with l = 10 m h and less than 0.5a with l = 5 m h. a low esr 22 m f, 25v solid tantalum capacitor (avx type tps or sprague type 593d) is adequate for mostapplications with the following caveat. solid tantalum capacitors can be destroyed with a very high turn-on surge current such as would be generated if a low impedance input source were ?ot switched?to the charger input. if this condition can occur, the input capacitor should have the highest possible voltage rating, at least twice the surge input voltage if possible. consult with the capacitor manufacturer before a final choice is made. a 4.7 m f ceramic capacitor such as the one used for the coupling capacitor can also be used.these capacitors do not have a turn-on surge limitation. the input capacitor must be connected directly to the v in pin and the ground plane close to the lt1513.output capacitor it is assumed as a worst case that all the switching output ripple current from the battery charger could flow in the output capacitor. this is a desirable situation if it is neces- sary to have very low switching ripple current in the battery itself. ferrite beads or line chokes are often inserted in series with the battery leads to eliminate high frequency currents that could create emi problems. this forces all the ripple current into the output capacitor. total rms current into the capacitor has a maximum value of about 1a, and this is handled with the two paralleled 22 m f, 25v capacitors shown in figure 1. these are avx type tps or sprague type 593dsurface mount solid tantalum units intended for switching applications. do not substitute other types without ensuring that they have adequate ripple current ratings. see input capacitor section for details of surge limitation on solid tantalum capacitors if the battery may be ?ot switched?to the output of the charger. coupling capacitor c2 in figure 1 is the coupling capacitor that allows a sepic converter topology to work with input voltages either higher or lower than the battery voltage. dc bias on the capacitor is equal to input voltage. rms ripple current in the coupling capacitor has a maximum value of about 1a at full charging current. a conservative formula to calculate this is: i ivv v coup rms chrg in bat in () () ( . ) () = + 11 2 (1.1 is a fudge factor to account for inductor ripple currentand other losses) with i chrg = 1.2a, v in = 15v and v bat = 8.2v, i coup = 1.02a. the recommended capacitor is a 4.7 m f ceramic type from marcon or tokin. these capacitors have extremely low esrand high ripple current ratings in a small package. solid tantalum units can be substituted if their ripple current rating is adequate, but typical values will increase to 22 m f or more to meet the ripple current requirements.diode selection the switching diode should be a schottky type to minimize both forward and reverse recovery losses. average diode current is the same as output charging current, so this will be under 2a. a 3a diode is recommended for most applications, although smaller devices could be used at reduced charging current. maximum diode reverse voltage will be equal to input voltage plus battery voltage. diode reverse leakage current will be of some concernduring charger shutdown. this leakage current is a direct drain on the battery when the charger is not powered. high kool m m is a registered trademark of magnetics, inc. metglas is a registered trademark of alliedsignal inc. downloaded from: http:///
9 lt1513/lt1513-2 sn1513 1513fas applicatio n s i n for m atio n wu u u current schottky diodes have relatively high leakage cur-rents (5 m a to 500 m a) even at room temperature. the latest very-low-forward devices have especially high leakage cur-rents. it has been noted that surface mount versions of some schottky diodes have as much as ten times the leakage of their through-hole counterparts. this may be because a low forward voltage process is used to reduce power dissipation in the surface mount package. in any case, check leakage specifications carefully before making a final choice for the switching diode. be aware that diode manufacturers want to specify a maximum leakage current that is ten times higher than the typical leakage. it is very difficult to get them to specify a low leakage current in high volume production. this is an on going problem for all battery charger circuits and most customers have to settle for a diode whose typical leakage is adequate, but theoretically has a worst-case condition of higher than desired battery drain. thermal considerations care should be taken to ensure that worst-case conditions do not cause excessive die temperatures. typical thermal resistance is 30 c/w for the r package but this number will c1,c3,c5 and r3tied directly to ground plane ground plane lt1513 tab and ground pin soldered to ground plane v bat c1 c1 d1 c5 r3 c3 2 winding inductor l1a l1b v in c2 lt1513 ?f04 r5 + ++ figure 4. lt1513 suggested partial layout for critical thermal and electrical paths vary depending on the mounting technique (copper area,airflow, etc.). average supply current (including driver current) is: im a vi v in bat chrg in =+ 4 0 024 () ( ) ( .) switch power dissipation is given by: p irvv v v sw chrg sw bat in bat in = + () () ( ) () () 2 2 r sw = output switch on resistance total power dissipation of the die is equal to supply currenttimes supply voltage, plus switch power: p d(total) = (i in )(v in ) + p sw for v in = 10v, v bat = 8.2v, i chrg = 1.2a, r sw = 0.3 w , i in = 4ma + 24ma = 28ma p sw = 0.64w p d = (10)(0.028) + 0.64 = 0.92w downloaded from: http:///
10 lt1513/lt1513-2 sn1513 1513fas applicatio n s i n for m atio n wu u u programmed charging currentlt1513-2 charging current can be programmed with a dc voltage source or equivalent pwm signal, as shown in figure 5. in constant-current mode, i fb acts as a virtual ground. the i set voltage across r5 is balanced by the voltage across r4 in the ratio r4/r5.charging current is given by: i vrri r charge iset fbvos = () (/) C 45 3 i fb input current is small and can normally be ignored, but i fb offset voltage must be considered if operating over a wide range of program currents. the voltage across r3 atmaximum charge current can be increased to reduce offset errors at lower charge currents. in figure 5, i set from 0v to 5v corresponds to an i charge of 0a to 1a +37/ 62ma. c4 and r4 smooth the switch current wave- form. during constant-current operation, the voltage feed- back network loads the fb pin, which is held at v ref by the i fb amplifier. it is recommended that this load does not exceed 60 m a to maintain a sharp constant voltage to constant current crossover characteristic. i charge can also be controlled by a pwm input. assuming the signal isa cmos rail-to-rail output with a source impedance of less than a few hundred ohms, effective i set is v cc multiplied by the pwm ratio. i charge has good linearity over the entire 0% to 100% range.voltage mode loop stability the lt1513 operates in constant-voltage mode during the final phase of charging lithium-ion and lead-acid batteries. this feedback loop is stabilized with a series resistor and capacitor on the v c pin of the chip. figure 6 shows the simplified model for the voltage loop. the error amplifier ismodeled as a transconductance stage with g m = 1500 m mho figure 6. constant-voltage small-signal model + r p ** 1m g m 1500 mho i p modulator section g m = = v in = dc input voltage v bat = dc battery voltage i p v1 4(v in ) v in + v bat v1 fb v c r1*71.5k r cap ? 0.15 each r bat 0.1 c1 c1 battery 1513 f06 c122 f each r212.5k 1.245v ea r g 330k r5330 * for 8.4v battery. adjust value of r1 for actual battery voltage ** r p and c p model phase delay in the modulator c50.1 f c p ** 3pf + + this is a simplified ac model for the lt1513 in constant-voltage mode. resistor and capacitor numbers correspond to those used in figure 1. r p and c p model the phase delay in the modulator. c3 is 3pf for a 10 h inductor. it should be scaled proportionally for other inductor values (6pf for 20 h). the modulator is a transconductance whose gain is a function of input and battery voltage as shown. as shown, this loop has a unity-gain frequency of about 250hz. unity-gain will move out to several kilohertz if battery resistance increases to several ohms. r5 is not used in all applications, but it gives better phase margin in constant-voltage mode with high battery resistance. figure 5 c40.1 m f r30.2 w 1513 f05 l1b i set r5 249k r4 10k i fb lt1513-2 downloaded from: http:///
11 lt1513/lt1513-2 sn1513 1513fas applicatio n s i n for m atio n wu u u figure 7. constant-current small-signal model problem, and indeed small signal loop stability can beexcellent even in the presence of subharmonic switching. the primary issue with subharmonics is the presence of emi at frequencies below 500khz. constant-current mode loop stability the lt1513 is normally very stable when operating in con- stant-current mode (see figure 7), but there are certain con- ditions which may create instabilities. the combination of higher value current sense resistors (low programmed charg- ing current), higher input voltages, and the addition of a loop compensation resistor (r5) on the v c pin may create an un- stable current mode loop. (a resistor is sometimes added inseries with c5 to improve loop phase margin when the loop is operating in voltage mode.) instability results because loop gain is too high in the 50khz to 150khz region where excess phase occurs in the current sensing amplifier and the modulator. the i fba amplifier (gain of ?2.5) has a pole at approximately 150khz. the modulator section con-sisting of the current comparator, the power switch and the magnetics, has a pole at approximately 50khz when the coupled inductor value is 10 m h. higher inductance will reduce the pole frequency proportionally. the design procedure pre-sented here is to roll off the loop to unity-gain at a frequency of 25khz or lower to avoid these excess phase regions. (from the electrical characteristics). amplifier output resis-tance is modeled with a 330k resistor. the power stage (modulator section) of the lt1513 is modeled as a transcon- ductance whose value is 4(v in )/(v in + v bat ). this is a very simplified model of the actual power stage, but it is sufficientwhen the unity-gain frequency of the loop is low compared to the switching frequency. the output filter capacitor model includes its esr (r cap ). a series resistance (r bat ) is also assigned to the battery model.analysis of this loop normally shows an extremely stable system for all conditions, even with 0 w for r5. the one condition which can cause reduced phase margin is with a very large battery resistance (> 5 w ), or with the battery replaced with a resistive load. the addition of r5 gives goodphase margin even under these unusual conditions. r5 should not be increased above 330 w without checking for two possible problems. the first is instability in the constantcurrent region (see constant-current mode loop stability), and the second is subharmonic switching where switch duty cycle varies from cycle to cycle. this duty cycle instability is caused by excess switching frequency ripple voltage on the v c pin. normally this ripple is very low because of the filtering effect of c5, but large values of r5 can allow highripple on the v c pin. normal loop analysis does not show this + + r p ** 1m g m 1500 mho i p modulator section i p = 4(v1)(v in ) v in + v bat v1 fb v c 1513 f07 1.245v ea i fba voltage gain = 12 r g 330k r5330 c50.1 f c a 10pf r a 100k r4 24 r30.1 c p 3pf c40.22 f i fb this is a simplified ac model for the lt1513 in constant-current mode. resistor and capacitor numbers correspond to those used in figure 1. r p and c p model the phase delay in the powerpath. c3 is 3pf for a 10 h inductor. it should be scaled proportionally for other inductor values (6pf for 20 h). the powerpath is a transconductance whose gain is a function of input and battery voltage as shown. the current amplifier has a fixed voltage gain of 12. its phase delay is modeled with r a and c a . the error amplifier has a transconductance of 1500 mho and an internal output shunt resistance of 330k.as shown, this loop has a unity-gain frequency of about 27khz. r5 is not used in all applications, but it gives better phase margin in constant voltage mode. downloaded from: http:///
12 lt1513/lt1513-2 sn1513 1513fas applicatio n s i n for m atio n wu u u the suggested way to control unity loop frequency is toincrease the filter time constant on the i fb pin (r4/c4 in figures 1 and 7). the filter resistor cannot be arbitrarilyincreased because high values will affect charging current accuracy. charging current will increase by 1% for each 40 w increase in r4. there is no inherent limitation on the value of c4, but if this capacitor is ceramic, it should be anx7r type to maintain its value over temperature. x7r dielectric requires a larger footprint. the formula for calculating the minimum value for the filter capacitor c4 is: c rv r fr v v in in bat 4 3 4 12 1500 5 24 = + ()()( )()( )() ()( )( ) m p v in = highest input voltage 1500 m = transconductance of error amplifier (ea)f = desired unity-gain frequencyv bat = battery voltage for example, assume v in(max) = 15v, r3 = 0.4 w (charging current set to 0.25a), r4 = 24 w , r5 = 330 w and v bat = 8v, cf 4 0 4 4 15 12 0 0015 330 15 8 1 = + = . ( )( )( )( . )( ) ) 6.3(25000)(39)( m the value for c4 could be reduced to a more manageable sizeby increasing r4 to 75 w and reducing r5 to 300 w , yielding 0.47 m f for c4. the 2% increase in charging current can be ignored or factored into the value for r3.more help linear technology field application engineers have a cad spreadsheet program for detailed calculations of circuit operating conditions. in addition, our applications depart- ment is always ready to lend a helping hand. the lt1371 data sheet may also be helpful. the lt1513 is identical except for the current amplifier circuitry. downloaded from: http:///
13 lt1513/lt1513-2 sn1513 1513fas typical applicatio n s u lithium-ion battery charger withswitchable charge current many battery chemistries require several constant-current settings during the charging cycle. the circuit shown in figure 8 uses the lt1513-2 to provide switchable 1.35a and 0.13a constant-current modes. the circuit is based on a standard sepic battery charger circuit set to a single lithium-ion cell charge voltage of 4.1v. the lt1513-2 has i fb referenced to ground allowing a simple resistor network toset the charging current values. in constant-current mode, the i fb error amplifier drives the fb pin, increasing charging current, until i r4 is balanced by i r5 . i ir i r charge r fbvos = ()()C 5 4 3 there are several ways to control i r5 including dac, pwm or resistor network as shown here. if the lithium cell requiresprecharging, q1 is turned on, setting a constant current of 0.13a. when charge voltage is reached, q1 is turned off, programming the full charge current of 1.35a. as the cell voltage approaches 4.1v, the voltage sensing network (r1, r2) starts driving the v fb pin, changing the lt1513-2 to constant-voltage mode. as charging current falls, the outputremains in constant-voltage mode for the remainder of the charging cycle. when charging is complete, the lt1513-2 can be shut down with the s/s pin. lt1513-2 l1a ctx10-4 l1b gnd i fb 1513 f08 v sw v fb 1 76 3 52 v c v in v in s/s r3330 w r5 36k r4 4.7k c50.1 m f 4 c40.22 m f c2 4.7 m f d1 mbrs330t3 c122 f 25v 2 + c325 f 25v r7910 q1 r610k 3.3v r30.25 w r178.7k 0.5% li-ionrechargable cell gnd r234k 0.5% + charge shutdown precharge charge figure 8. lithium-ion battery charger downloaded from: http:///
14 lt1513/lt1513-2 sn1513 1513fas typical applicatio n s u this cold cathode fluorescent lamp driver uses a royerclass self-oscillating sine wave converter to driver a high voltage lamp with an ac waveform. ccfl royer converters have significantly degraded efficiency if they must operate at low input voltages, and this circuit was designed to handle input voltages as low as 2.7v. therefore, the lt1513 is connected to generate a negative current through l2 that allows the royer to operate as if it were connected to a constant higher voltage input. the royer output winding and the bulb are allowed to float inthis circuit. this can yield significantly higher efficiency in situations where the stray bulb capacitance to surrounding enclosure is high. to regulate bulb current in figure 9, royer input current is sensed with r2 and filtered with r3 and c6. this negative feedback signal is applied to the i fb pin of the lt1513. for more information on this circuit contact the ltcapplications department and see design note 133. consid- erable written application literature on royer ccfl circuits is also available from other ltc application and design notes. 2.7v to 20v l1 20 m h c24.7 m f ceramic c1 47 m f elect r4 20k r3 10k r5330k pwm dimming ( ? 1khz) 5 1 2 3 10 6 t1 4 c54.7nf c91 m f c10 10nf d13a r1470 w c3 0.082 m f wima c427pf 3kv lamp current 5.6ma negative voltageis generated here r2 0.25 w 1513 f09 r72k d2 15v 3 2 4, tab 7 6 1 5 l2 20 m h ccfl c60.1 m f + q2 q1 c2: tokin multilayer ceramicc3: must be a low loss capacitor, wima mkp-20 or equivalent l1, l2: coiltronics ctx20-4 (must be separate inductors) q1, q2: zetex ztx849 or fzt849 t1: coiltronics ctx110605 (67:1) lt1513-2 v fb v in v sw i fb gnd s/s v c figure 9. ccfl power supply for floaing lamp configuration operates on 2.7v downloaded from: http:///
15 lt1513/lt1513-2 sn1513 1513fas package descriptio n u dimensions in inches (millimeters) unless otherwise noted. information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen-tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. r package 7-lead plastic dd pak (ltc dwg # 05-08-1462) r (dd7) 0396 0.026 ?0.036 (0.660 ?0.914) 0.143 +0.012 0.020 () 3.632 +0.305 0.508 0.040 ?0.060 (1.016 ?1.524) 0.013 ?0.023 (0.330 ?0.584) 0.095 ?0.115 (2.413 ?2.921) 0.004 +0.008 0.004 () 0.102 +0.203 0.102 0.050 0.012 (1.270 0.305) 0.059 (1.499) typ 0.045 ?0.055 (1.143 ?1.397) 0.165 ?0.180 (4.191 ?4.572) 0.330 ?0.370 (8.382 ?9.398) 0.060 (1.524) typ 0.390 ?0.415 (9.906 ?10.541) 15 typ 0.300 (7.620) 0.075 (1.905) 0.183 (4.648) 0.060 (1.524) 0.060 (1.524) 0.256 (6.502) bottom view of dd pak hatched area is solder plated copper heat sink t7 package 7-lead plastic to-220 (standard) (ltc dwg # 05-08-1422) 0.040 ?0.060 (1.016 ?1.524) 0.026 ?0.036 (0.660 ?0.914) t7 ( to-220 ) ( formed ) 1197 0.135 ?0.165 (3.429 ?4.191) 0.700 ?0.728 (17.780 ?18.491) 0.045 ?0.055 (1.143 ?1.397) 0.165 ?0.180 (4.191 ?4.572) 0.095 ?0.115 (2.413 ?2.921) 0.013 ?0.023 (0.330 ?0.584) 0.620 (15.75) typ 0.155 ?0.195 (3.937 ?4.953) 0.152 ?0.202 (3.860 ?5.130) 0.260 ?0.320 (6.604 ?8.128) 0.147 ?0.155 (3.734 ?3.937) dia 0.390 ?0.415 (9.906 ?10.541) 0.330 ?0.370 (8.382 ?9.398) 0.460 ?0.500 (11.684 ?12.700) 0.570 ?0.620 (14.478 ?15.748) 0.230 ?0.270 (5.842 ?6.858) downloaded from: http:///
16 lt1513/lt1513-2 sn1513 1513fas ? linear technology corporation 1996 linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 fax: (408) 434-0507 telex: 499-3977 www.linear-tech.com lt/tp 0198 rev a 4k ? printed in the usa part number description comments lt1239 backup battery management system charges backup battery and regulates backup battery output when main battery removed ltc 1325 microprocessor controlled battery management system can charge, discharge and gas gauge nicd, nimh and pb-acid batteries with software charging profiles lt1510 1.5a constant-current/constant-voltage battery charger step-down charger for li-ion, nicd and nimh lt1511 3.0a constant-current/constant-voltage battery charger step-down charger that allows charging during computer operation an d with input current limiting prevents wall-adapter overload lt1512 sepic constant-current/constant-voltage battery charger step-up/step-down charger for up to 1a current related parts downloaded from: http:///

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